RF and Microwave transistor chips are best characterized “on wafer”. This allows avoidance of parasitic connection elements, like wire bonds and fringe capacitors, which are associated with packaging the devices in order to mount them in test fixtures. It also allows a much larger number of devices to be tested “in situ” without having to laboriously slice the wafer, mount and wire-bond each individual chip. The “on wafer” testing is at this time the preferred testing method, except for very high power devices, beyond 10 Watt RF power. On-wafer testing is also the exclusive testing method in millimeterwave frequencies, since device packaging is extremely difficult and the parasitic elements associated with the package (inductance of wire bonds and fringe capacitors of package housings) would falsify the measured data to the point of uselessness.
However, on-wafer testing implies reduced tuning range. This is because of the insertion loss of the connection between the tuner and the wafer probes (11) and the insertion loss of the probes themselves (FIG. 1). The DUT, including wafer probes or embedded in a test fixture, is represented schematically in box (12). To compensate for the said insertion loss active load pull has been introduced (see ref. 1). The latest comprehensive active load pull technique is the so called “active injection” load pull (FIG. 1). In this setup a harmonic receiver (13) is used which also provides two or more coherent signal sources (14, 15); coherent means the signals have a fixed controlled phase between them. In this configuration signal (15) being injected into the output (11) of the DUT is at the same frequency and has its phase controlled electronically relative to the phase of signal (14). The signal (14) is injected into the input of the DUT through an (optional) driver amplifier (not shown) and an impedance tuner (16) and an input directional coupler (17). The coupler (17) feeds (18) a small part of the injected (a1) and reflected (b1) signal waves into the harmonic receiver (13). The signal going out of the DUT (11) passes through another directional coupler (19), which also feeds a small part of the injected (a2) and reflected (b2) signal waves into the harmonic receiver (13), and said outgoing signal (11) interacts with the signal injected from the power amplifier (111). Through amplitude and phase control of the injected signal (15) a virtual reflection factor r is generated at the output (11) of the DUT (12): r=a2/b2. Since the amplifier (111) can increase the injected signal amplitude at will, the power returning from the load (a2) can be made larger than the power leaving the DUT (b2). In practical terms this means a reflection factor r>1. This allows matching any internal impedance (reflection factor) of the DUT despite any insertion losses.
The relation between reflection factor and impedance is: Z=Zo*(1+Γ)/(1−Γ); where Zo is a standard impedance (typically 50 Ohm); this means that for r=−1, Z=O Ohm.
The actual problem with this configuration is that there exist, typically, a large impedance mismatch between the output of the power amplifier (111) and the output of the DUT (11). Typical impedances of power transistors (DUT) is 0.5-2 Ohms. Typical impedance of power amplifiers is 50 Ohms. This creates a large mismatch ratio between 25:1 and 100:1. The consequence is that the power required from the amplifier (111) is typically 20 times larger or more than the power generated by the DUT (see ref. 4). This requirement can be reduced if a transformer is used between the DUT (12) and the amplifier (111). This is shown in FIG. 2. The impedance Tuner (21) transforms the (typically) 50 Ohm output impedance of the amplifier (22, 111) closer to the (typically 0.5-2 Ohms) output impedance of the DUT (23). The mismatch reduction (improvement) factor can reach high values, depending on the impedance the tuner (21) can generate and present to the DUT (23). The mismatch factor can be expressed as a “standing wave voltage ratio VSWR”; in a system with a characteristic impedance of 50 Ohm VSWR=50 Q/R.dut, where R.dut is the internal output impedance of the DUT (typically 0.5-2 Ohms). This gives an idea of the actual mismatch: VSWR is between 25 and 100. A transformer (such a tuner) will create, typically, a pre-match factor of the order of 5:1 to 15:1. Knowing that the total mismatch is the product of the partial mismatches, in this case the total mismatch will be reduced by this factor; a 25:1 initial mismatch will be reduced by the said tuner/transformer to values between 5:1 and 1.7:1 (=25/15:1) and an initial mismatch of 100:1 will be reduced to values between 20:1 and 6.7:1. If the tuner can actually reach exactly the conjugate impedance of the DUT (tuner VSWR=DUT VSWR) then no additional signal power (24) will be needed (see ref. 4). If it can only reach values close to the conjugate internal impedance of the DUT then only a reduced signal power (24) is required (see ref. 2). This reduces significantly the complexity and especially the cost of the test setup, since power amplifiers can be expensive.
Harmonic tuning, i.e. independent impedance control at the harmonic frequencies can be materialized using two or more signals (31, 32) injected into a frequency combiner (35) and then into the output of the DUT (FIGS. 3 and 7). These signals must be coherent with the input signal at frequency Fo (34), whereas one signal (31) is at the same, fundamental, frequency (Fo) and the other (32) is typically at the first or even at a higher harmonic frequency (2 Fo or 3 Fo etc. . . . ). The problem with the setup of FIG. 3 is that a wideband impedance tuner (33) is used. A wideband tuner creates reflection over a large frequency range, which, typically, includes the fundamental and several harmonic frequencies. Whereas controllable impedances are created only at one (typically the fundamental, frequency Fo) uncontrollable reflections are created at any other frequency. It may therefore happen that the reflection factors shown to the DUT at a harmonic frequency is slightly or radically different than the internal impedance of the DUT at this frequency (FIG. 4). The dots in FIG. 4 show a scenario where the uncontrollable reflections at the harmonic frequencies (2 Fo and 3 Fo) are anti-diametric (180 degrees off) to the optimum reflection factors of the DUT at said frequencies. This is not the usual scenario, but a possible one, that an efficient test setup should be able to handle routinely.
In FIG. 4 the dot marked “Fo” is noted as “reduced because of coupler loss”: this means that a typical setup as discussed here (FIGS. 1, 2, 3) includes directional couplers adjacent to the DUT (17, 19) in order to detect the incident and reflected waves by the receiver (13). In such a case the insertion loss in said couplers will reduce the tuning range of said tuner (21). This explains the markings on “Fo” in FIG. 4.
The effect of harmonic impedance mismatch (FIG. 4) is shown quantitatively in FIGS. 5 and 6: the plot in FIG. 5 represents a calculation of injected power required to match the harmonic impedance of a DUT as a function of the phase mismatch between the reflection factor generated by the wideband tuner (33) and the DUT. The actual numbers shown are for a typical 20 Watt DUT operated at medium compression and generating harmonic power at 2 Fo of 0.4 Watt (P(2 Fo)/P(Fo)=−17 dB). The plot is typical for many practical cases. It shows at point (52) that, if the mismatch angle is 180 degrees the requirement for harmonic injection power increases by a factor of 33.75 (13.5/0.4) relative to the case where proper pre-matching reduces the injection power requirement is minimum or even zero, as shown at point (51). A similar calculation holds for 3 Fo as well. Because harmonic frequencies are high, associated power amplifiers are very expensive (see ref. 3). This is all in addition to any unavoidable connection and adapter losses. Therefore an appropriate means for reducing said power requirement will reduce sensibly the cost of the setup.